FIG. 1 is a reduced complexity circuit diagram of a typical synchronous buck mode DC-DC converter architecture for charging a battery by way of a voltage that is supplied to the charger circuitry and to downstream powered circuitry from an AC-DC adapter. As shown therein, a powered system bus 10 is coupled to a system power source such as an AC-DC adapter, which is operative to supply a prescribed DC voltage, such as a voltage value on the order of sixteen to nineteen volts DC, that is to be available for powering one or more system bus loads 12, which are connected between the powered bus 10 and a reference voltage bus 14, such as a zero volts or ground bus. In addition to supplying a DC voltage to system bus components, the system bus is employed to charge an auxiliary power storage device, such as a battery 16.
For this purpose, an upper controlled switch or MOSFET 21 and a lower controlled switch or MOSFET 23 have their source-drain paths coupled in series between the system bus 10 and the reference voltage bus 14. The gates of these two MOSFETs are adapted to be driven by respective (complementary) pulse width modulation (PWM) signals supplied thereto by a PWM controller. The common or phase node 25 between the upper MOSFET or UFET 21 and the lower MOSFET or LFET 23 is coupled by way of an inductor 27 to an output node 29 to which the battery 16, referenced to the ground bus 14, is coupled. In addition, a capacitor 33 is coupled between output node 29 and the ground reference bus 14.
Now although the use of a synchronous buck mode DC-DC converter architecture provides a relatively efficient mechanism for charging the battery, its operation can lead to the delivery of a negative or reverse current from the battery charger onto the system supply bus 10, thereby increasing the system bus voltage to unsafe levels that may damage downstream system components. Such a flow of negative current can result from a number of events, such as, but not limited to, soft starting the charger, inserting the battery, and removing the adapter voltage. In these events, the charger is operating open loop with a duty cycle that is lower than the closed loop duty cycle. It is possible to boost the system bus, if negative inductor current is flowing, namely, away from the battery opposite the direction of the arrow A, which shows the direction of positive inductor current flow into the battery, and when the system bus load is low (i.e., the powered system, such as a laptop computer is off and the battery is being charged). Current boosting into the system bus cannot go into the AC-DC adapter (as it is not designed to sink current), or be used by the load (which is turned off), so that the system bus voltage rises.
The mechanism through which negative current makes its way to the system bus is as follows. When the UFET 21 is turned off and the LFET 23 is turned on, the current in inductor 27 will decrease to zero and then become negative, and current will begin to flow from the battery through the inductor 27 in the negative direction, and down through the LFET to the return bus or ground. This is the current loop through which current will flow when the LFET 23 is on. When the LFET is turned off, the current that has built up in the inductor 27 cannot go through the LFET, and instead flows through the body diode of the UFET 21 to the supply bus 10, thereby undesirably boosting the supply bus voltage, typically by a value on the order of several or more volts—high enough to damage loads connected to the system bus.
To address this problem, designers of synchronous buck mode DC-DC converters have commonly employed a mechanism, known as diode emulation, which causes the LFET to behave as though it were a diode. In this diode emulation mode, the direction of current flow through the LFET is monitored. As long as current is flowing in the positive direction (from the source to the drain) the LFET 23 is allowed to be turned on. However, if the current reaches zero or goes negative, then the lower FET is turned off. This effectively makes the lower FET emulate a diode, in that the LFET allows positive current to flow through it (upwardly from the source to the drain and out through the inductor in the positive direction), but blocks current in the opposite or negative direction, in that no current is allowed to flow through the LFET in the drain-to-source direction, once the current reaches a zero value.
A reduced complexity schematic of a conventional circuit for implementing this diode-emulation control function is shown diagrammatically in FIG. 2, as comprising a phase comparator 40, having its positive or non-inverting (+) input 41 coupled to the drain and its negative or inverting (−) input 42 coupled to the source of LFET 23. The output of the phase comparator 40 is coupled to one input of a NOR gate 45, a second input of which is coupled to receive the PWM signal. The output of NOR gate 45 is coupled through a driver 46 to the gate input of LFET 23. Similarly, the PWM signal is coupled through a driver 47 to the gate input of UFET 21, and further to a delay circuit 50, the output of which is coupled to the disable input of phase comparator 40. Delay circuit 50 is used to disable or ‘blank’ the operation of phase comparator 40 for a prescribed time delay (e.g., on the order of 200 ns) subsequent to the rising edge of the PWM signal, to allow ringing at the phase node 25 associated with the inductance of inductor 27 and the parasitic capacitance of the phase node 25 to subside sufficiently to allow an accurate measurement of current flow.
The operation of the circuit of FIG. 2 may be understood with reference to the set of waveforms shown in FIG. 3. When the PWM waveform shown at 300, transitions from high to low at time 301, the voltage at the phase node 25, which had previously been at Vin due to the conduction of UFET 21, will undergo negative ringing below zero volts as shown by the ringing portion 311 of phase node voltage waveform 310. Because the ringing associated with the PWM transition constitutes noise, the operation of the phase comparator 40 is blanked by the delay circuit 50 for a period of time that allows the ringing to subside. At the end of the ringing interval shown at 312, the phase node voltage is negative and begins a gradual transition towards zero volts as the inductor current gradually transitions towards zero as shown at 313. At this point, the inductor current can be validly measured.
A voltage representative of the inductor current is produced by the on-resistance of the LFET 23 and value of the negative inductor current flowing from the drain to the source of LFET 23. Because the source of LFET 23 is connected to ground, then when the current is positive—flowing from source to drain—the voltage at the phase node is actually below ground, as shown at 313, referenced above. Once the voltage at the phase node has increased to zero volts, at time 314, the output of the phase comparator 40 changes state and, via NOR gate 45, turns off the LFET 23, so that the LFET will act as a diode for negative inductor current.
The waveforms of FIG. 4 illustrate a fundamental problem with the mechanism employed in the circuit of FIG. 3. If diode emulation were not employed, then the PWM signal for controlling the turn on/off of the LFET 23 would be the complement of the PWM signal employed for the UFET 21. However, since diode emulation is controlled by the presence of the delay circuit 50, the NOR gate 45, and the phase comparator 40, the LFET 23 has a shorter on time than the inverse of PWM waveform applied to the gate of UFET 21. The top waveform 400 of FIG. 4 corresponds to the PWM signal that is applied to the gate UG of the UFET 21, while the bottom waveform 420 corresponds to the PWM signal that is applied to the gate LG of the LFET 23. The intermediate waveform 410 in FIG. 4 represents the variation in the inductor current through inductor 27.
As shown in FIG. 4, in response to the rising edge 401 in the PWM waveform 400 applied to the gate UG of the UFET 21, inductor current begins a positive ramp at 411, until the high-to-low transition 402 in the PWM waveform 400. In response to this transition, the UFET 21 is turned off, and the inductor current begins ramping down towards 0 amps, as shown at 412. In addition, when the PWM waveform applied to gate of the UFET 21 goes low, the waveform 420 applied to the gate of the LFET 23 goes high at 421, thereby turning on the LFET 23. During the interval between the high-to-low transition 402 in the PWM waveform applied to gate UG of the UFET 21 and the time 413 at which the inductor current reaches zero, positive inductor current is being supplied by the LFET 23, which has been turned on by the low-to-high 421 transition in the LG PWM signal 420.
The positive inductor current being supplied by LFET 23 flows from its source, which is at ground potential, to its drain, which is at a phase node voltage negative with respect to ground. When the inductor current reaches zero amps (0 A) at time 413, one would like to turn off the LFET 23. However, due to the use of the delay/blanking interval 414, the inductor current is not being monitored, so that no turn off signal is applied to the gate of the LFET 23. Instead, the inductor current continues to decrease well below zero amps, as shown at 415. Finally, at the end of the blanking interval, the output of the phase comparator 40, which has detected that Vd>Vs, is allowed to indicate that negative inductor current has been detected, and the LFET 23 is turned off. This is shown in FIG. 4 by the high-to-low transition 422 of the gate control waveform LG 420 to the gate input of the LFET 23.
When the LFET 23 turns off, the phase node 25 will go from zero volts to a diode drop above Vin, so that the body diode of UFET 21 is conducting. With both UFET 21 and LFET 23 now turned off, the negative polarity inductor current begins to ramp up towards zero amps, as shown at 416. During this transition, the negative inductor current is flowing through the body diode of the UFET 21. Eventually, at 417, the ramping up negative current reaches zero amps and the cycle described above repeats.
An examination of the inductor current waveform 410 reveals that the average inductor current is negative, as shown by broken lines 418. This means that an average negative current is being supplied by the battery into the system bus—placing the system bus 10 at an undesirably high voltage value. It will be readily appreciated, therefore, that within the blanking interval 414 a fairly large negative inductor current is realized. If the battery voltage is relatively high and the value L of the inductor 27 is relatively low, then di/dt is relatively large; namely, the inductor current reaches a relatively large negative value within a relatively small window of time. One way to mitigate against this effect is to reduce the blanking interval. However, doing so creates the risk that the phase comparator will trigger on a ringing edge rather than on a true zero-crossing ramp, as described above with reference to FIG. 3. As pointed out above, the ringing is due to the parasitic capacitance of the phase node and the value of the inductance. The blanking interval must be kept sufficiently wide to allow the ringing voltage at the phase node to subside. However, doing so means that there will be a fairly substantial average negative inductor current presented to the system bus, which is the problem to be solved.